Composite amplifier with optimized linearity and efficiency

ABSTRACT

The invention relates to a composite amplifier ( 500 ) based on a main amplifier ( 510 ) and an auxiliary amplifier ( 520 ), and compensation for non-linear amplifier behavior by means of respective non-linear models ( 570, 575 ) of parasitic. In order to provide proper excitation of the non-linear models. a filter network ( 560 ) based on a linear model of the output network of the composite amplifier is provided. The linear filter network ( 550 ) basically determines ideal output node voltages, which are used as input to the respective non-linear models for generating appropriate compensation signals. The compensation signals are finally merged into the input signals of the respective sub-amplifiers ( 510, 520 ), thus effectively compensating for the effects of the non-linear parasitics.

TECHNICAL FIELD OF THE INVENTION

[0001] The present invention generally relates to composite amplifiersand more particularly to techniques for optimizing the linearity as wellas efficiency of such amplifiers.

BACKGROUND OF THE INVENTION

[0002] In cellular base stations, satellite communication systems aswell as other communication and broadcasting systems of today, it isoften desirable to amplify multiple radio frequency (RF) channelssimultaneously in the same power amplifier instead of using a dedicatedpower amplifier for each channel. However, when using one and the samepower amplifier for the simultaneous amplification of multiple RFchannels spread across a fairly wide bandwidth, a high degree oflinearity is required so that the phases and amplitudes of all thesignal components are preserved in the amplification process.

[0003] If the linearity is inadequate, the simultaneously amplifiedchannels cross-modulate, causing interference in these and otherchannels. The non-linearities manifest themselves as cross-modulation ofdifferent components of the signal, leading to leakage of signal energyto undesired channels. In addition, the spectra of the signal componentsare normally broadened, causing additional interference within thechannels or in other channels.

[0004] In addition to linearity, one of the most important properties ofa power amplifier is efficiency. The efficiency must be kept high inorder to reduce the need for cooling as well as the overall powerconsumption, and to increase the lifetime of the amplifier.

[0005] Consequently, the problem of enhancing the linearity must besolved without sacrificing the amplifier efficiency.

[0006] A common way of increasing the efficiency of an RF poweramplifier is to use the Doherty principle as described and developed inreferences [1-7]. FIG. 1 is a schematic block diagram of a conventionalDoherty amplifier. The Doherty amplifier 100 is a so-called compositeamplifier, which in its basic form comprises two sub-amplifier stages, amain amplifier 110 and an auxiliary amplifier 120. The auxiliaryamplifier 120 is connected directly to the load 130, and the mainamplifier 110 is connected to the load through an impedance inverter140, usually in the form of a quarter wavelength transmission line or anequivalent lumped network

[0007] At low output levels, only the main amplifier 110 is active. Inthis region, the main amplifier 110 sees a higher (transformed) loadimpedance than the impedance at peak power, which results in increasedefficiency. The input drive arrangement 150 of the auxiliary amplifier120 is configured with a non-linear drive function f2(x) such that whenthe output level climbs over the so-called transition point (usuallyhalf the maximum output voltage), the auxiliary amplifier kicks in,starting to drive current into the load 130. Through theimpedance-inverting action of the quarter wave transmission line 140,the effective impedance at the output of the main amplifier 110 isreduced such that the main amplifier is kept at a constant maximumvoltage above the transition point. The key action of the Dohertyamplifier occurs in the region where the auxiliary amplifier 120 isactive, and the main amplifier 110 is close to its maximum voltagecondition, with high overall efficiency as a result.

[0008] However, conventional Doherty amplifiers only providesatisfactory linear performance and efficiency in a relatively narrowfrequency band. The quarter wavelength impedance inverter provides aphase shift of 90 degrees only at a single frequency, and the output ofa practical Doherty amplifier will be distorted at frequencies away fromthis so-called center frequency because of a reflection of the outputcurrent of the auxiliary amplifier at the impedance inverter. Losses inthe transistors, the impedance inverter and the DC feed networks mayalso give rise to unexpected distortion. In addition to these sources ofdistortion, Doherty amplifiers will in practice always suffer fromnon-linearities caused by non-linear output parasitic elements such asparasitic conductances and capacitances, commonly referred to asparasitics.

[0009] It is generally known that the non-linearities encountered inDoherty amplifiers are strongly frequency-dependent. The non-linearitieswill manifest themselves both as (modulated) harmonic overtones andintermodulation products. The intermodulation products are the mostsevere for communication systems since the harmonic overtones can befiltered away before the signal reaches the antenna. The intermodulationproducts on the other hand appear right among the desired signals andcan generally not be filtered away before transmission. The complexfrequency dependency makes it very difficult to compensate for thenon-linear intermodulation products by using pre-distortion. Simplepre-distortion techniques can not compensate for these non-linearities.In fact, a very complex and hence expensive pre-distorter implementedwith digital signal processing (DSP) techniques will be required. Such acomplex pre-distorter is furthermore difficult to adjust properly andwill generally not optimize the efficiency.

[0010] Consequently, there is a general demand for an improved techniqueof compensating for non-linearities in a composite amplifier.

RELATED ART

[0011] Reference [8] discloses a circuit technique for cancelingnon-linear capacitor-induced harmonic distortion in a single amplifier.The amplifier transistor is associated with a non-linear capacitancethat produces an undesirable non-linear current. An additionalcompensating transistor that has a similar non-linear capacitance isused together with a current mirror to produce a correction current thatcancels the undesirable non-linear current.

SUMMARY OF THE INVENTION

[0012] It is a general object of the present invention to improve thelinearity of a composite amplifier.

[0013] It is a particular object of the invention to compensate fornon-linearities in a composite amplifier, especially those caused byparasitics in the amplifier.

[0014] These and other objects are met by the invention as defined bythe accompanying patent claims.

[0015] The general idea according to the invention is to emulate andcompensate for a non-linear amplifier behavior based on a non-linearmodel of a parasitic, using a linear model of the output network of thecomposite amplifier to provide proper excitation of the non-linearmodel.

[0016] A careful analysis of the composite amplifier and the involvedparasitics reveals that the parasitics are highly dependent on theoutput node voltages of the sub-amplifiers within the compositeamplifier and that “ideal” output node voltages can be determined basedon a linear model of the output network of the composite amplifier. Bydetermining the output node voltages and using them as input torespective models of non-linear, voltage-dependent parasitics,appropriate compensation signals are emulated. The emulated compensationsignals are then merged into the input signals of the respectivesub-amplifiers, thus effectively compensating for the effects of thenon-linear parasitics.

[0017] It should be understood that the invention is not restricted toparasitics, but can be used to compensate for any non-linear amplifierbehavior that can be modeled as a non-linear parasitic.

[0018] It has also been shown that the non-linear output current of theauxiliary sub-amplifier in a Doherty-type composite amplifier not onlygenerates a desired voltage at the main sub-amplifier, but also ends upas distortion in the output of the composite amplifier, mainly due tothe reflection of the non-linear output current of the auxiliaryamplifier at the impedance inverter. In many cases, this distortion ismuch more severe than the distortion caused by non-linear parasitics.

[0019] Therefore, according to a preferred embodiment of the invention,the proposed compensation technique based on non-linear modeling ofparasitics is combined with a technique for linearly compensating forthe distortion that the non-linear output current of the auxiliaryamplifier causes at the output. Preferably, the non-linear outputcurrent from the auxiliary sub-amplifier is emulated based on a linearoutput network model and merged into the input signal of the mainsub-amplifier, thus effectively compensating for this large-scale typeof distortion caused by the reflection of the non-linear output currentat the impedance inverter.

[0020] The invention offers the following advantages:

[0021] Enhanced linearity, without sacrificing amplifier efficiency;

[0022] Compensation for complicated frequency-dependent non-linearities;and

[0023] Simple and effective implementation.

[0024] Other advantages offered by the present invention will beappreciated upon reading of the below description of the embodiments ofthe invention.

BRIEF DESCRIPTION OF THE DRAWINGS

[0025] The invention, together with further objects and advantagesthereof will be best understood by reference to the followingdescription taken together with the accompanying drawings, in which:

[0026]FIG. 1 is a schematic block diagram of a conventional Dohertyamplifier;

[0027]FIG. 2 is a schematic high-level block diagram of a radiotransmitter based on a composite power amplifier,

[0028]FIG. 3 is a schematic high-frequency model of a field-effecttransistor (FET), including parasitic elements;

[0029]FIG. 4 illustrates a linear model of the output network of aDoherty-type composite amplifier,

[0030]FIG. 5 is a schematic block diagram of a composite power amplifierwith non-linear compensation for parasitics according to an illustrativeembodiment of the invention;

[0031]FIG. 6 is a schematic block diagram of a composite power amplifierwith non-linear compensation for parasitics in combination withcompensation for the non-linear output current of the auxiliaryamplifier according to an illustrative embodiment of the invention; and

[0032]FIG. 7 is a schematic block diagram of a favorable, simplifiedimplementation of a composite amplifier according to the currently mostpreferred embodiment of the invention.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

[0033] Composite amplifiers can be found in various applications in manyfields of technology such as consumer electronics, radar technology andradio communication. In the following, the invention will be describedwith reference to a particular application within the field of radiocommunication. It should though be understood that the invention is notlimited thereto, and that other applications are feasible as well.

[0034] In a typical radio application, as schematically illustrated inthe high-level block diagram of FIG. 2, a composite power amplifier isarranged in a radio transmitter for simultaneous amplification ofseveral narrow-band channels. In a very basic realization, thetransmitter 200 comprises a general input unit 210 for combining theinput signals into a complex multi-channel signal, a composite poweramplifier (PA) 220 for simultaneous amplification of the multiplechannels and a transmission element 230. Such a basic realization ofcourse requires that the input signals are modulated RF signals withinthe desired target frequency band. If the input signals are base-bandsignals, up-conversion to the radio frequency band is also required. Thetransmitter illustrated in FIG. 2 is adapted for amplification andtransmission of for example several FDMA/TDMA carrier waves, but caneasily be modified for amplification and transmission of a carrier waveon which several CDMA channels are superimposed, or for multi-levellinear modulation formats.

[0035] In order to preserve the phases and amplitudes of all the signalcomponents in the amplification process and to prevent leakage ofinterfering signal energy between the channels, a high degree oflinearity is required in the composite amplifier. In this respect, ithas turned out to be particularly difficult to eliminate the effects ofnon-linear parasitics in the composite amplifier.

[0036] In general, non-linear parasitics in a composite amplifier end upas complex frequency-dependent non-linearities in the output that cannot be compensated for by simple pre-distortion techniques. The problemis mainly caused by the fact that some transistor node voltages,especially the drain (FET implementation) or collector (BJTimplementation) voltages of the involved transistors, are required to benon-linear by a desired non-linear shaping in order to maximizeefficiency even when the composite amplifier output should be perfectlylinear. It has been shown that the parasitics are highly dependent onthese output node voltages, and that the interaction of these non-linearnode voltages with the non-linear parasitics in a composite amplifierwill cause strongly frequency-dependent non-linearities in the output aswell as sub-optimal efficiency. This complex frequency dependence makesit very difficult to compensate for the non-linearities by usingpre-distortion.

[0037]FIG. 3 is a schematic high-frequency model of a field-effecttransistor (FET), including parasitic elements. The transistor 300 is aconventional FET with gate, source and drain terminals. The mostimportant parasitics are generally the drain-source capacitance C_(DS)and the drain-source resistance R_(DS), which cause non-linearities witha complicated frequency dependence in the output when they interact withthe drain-source output voltage V_(DS).

[0038] Since the parasitics, as discussed above, are dependent on theoutput node voltages of the sub-amplifiers within the compositeamplifier, it is desirable to analyze the voltage behavior at thesenodes. For this purpose, a linear model of the output network of thecomposite amplifier is used.

[0039]FIG. 4 illustrates a linear model of the output network of aDoherty-type composite amplifier. In this model of the output network400, the active part of the transistor output of each of thesub-amplifiers 410, 420 is modeled as linear controlled currentgenerator. The finite output conductances of the transistors, togetherwith possible reactances, are lumped together as z_(p1) and z_(p2),respectively. The impedances presented to the current generator outputnodes are defined as: $\begin{matrix}\begin{matrix}{{z_{11} = \frac{v_{1}}{i_{1}}}}_{i_{2} = 0} & {{z_{22} = \frac{v_{2}}{i_{2}}}}_{i_{1} = 0}\end{matrix} & (1)\end{matrix}$

[0040] Similarly, the transimpedances (the voltage at the inactiveamplifier output in response to an output current at the activeamplifier) are defined as: $\begin{matrix}\begin{matrix}{{z_{21} = \frac{v_{2}}{i_{1}}}}_{i_{2} = 0} & {{z_{12} = \frac{v_{1}}{i_{2}}}}_{i_{1} = 0}\end{matrix} & (2)\end{matrix}$

[0041] Assuming that all components are reasonably linear, superpositioncan be used for analyzing this model. The composite amplifier outputvoltage at the antenna 430 is here assumed to be the same as the outputvoltage at auxiliary amplifier 420, although in reality there can be afeeder cable, filters, etc. separating the actual antenna and theamplifier output. The combined effect of all these elements is includedin the antenna (output) impedance, z_(ANT). The transistor feedbackimpedances (mainly due to gate-drain capacitances) are neglected. Thisis perfectly valid if the feedback is small.

[0042] Based on the presented linear model of the output network, theoutput node voltages can be determined as:

v ₁ =z ₁₁ ·i ₁ +z ₁₂ ·i ₂   (3)

v ₂ =z ₂₂ ·i ₂ +z ₂₁ ·i ₁   (4)

[0043] By applying the determined output node voltages to respectivemodels of non-linear voltage-dependent output parasitics, the non-linearbehavior of the parasitics in the corresponding outputs can be emulated.By merging the emulated output behavior of the parasitics into therespective input signals to the sub-amplifiers, the effects of thenon-linear output parasitics can be compensated for.

[0044]FIG. 5 is a schematic block diagram of a composite power amplifierwith non-lincar compensation for parasitics according to an illustrativeembodiment of the invention. The composite amplifier 500 basicallycomprises a main amplifier 510, an auxiliary amplifier 520, a load 530in the form of an antenna, an impedance inverter 540, and an input drivearrangement 550 for the auxiliary amplifier. In order to emulate thenon-linear behavior of output parasitics present in the compositeamplifier, a filter network 560 is introduced together with respectivenon-linear parasitic models 570, 575.

[0045] In this embodiment, the filter network 560 is implementedaccording to expressions (3)(4) above with the filters G₁₁-G₂₂ emulatingthe impedances and transimpedances of the output network of FIG. 4:

G₁₁=z₁₁  (5)

G₁₂=z₁₂   (6)

G₂₁=z₂₁   (7)

G₂₂=z₂₂   (8)

[0046] The filter network 560 basically determines the “ideal” (withoutnon-linear parasitics) transistor output node voltages. The determinedoutput node voltages are then applied to the non-linear models 570 and575, respectively, to emulate the currents through the parasitics. Byusing the emulated currents as compensation currents and adding them tothe original input currents in the adder elements 572, 577, the effectsof the non-linear parasitics are effectively cancelled. In order tosynchronize of the original current signals with the emulatedcompensation currents, delay elements 580, 585 are used.

[0047] This technique offers a relatively simple way of reducingcomplicated frequency-dependent non-linearities due to non-linearparasitics in composite amplifiers. Since the technique is based onthorough knowledge of how composite amplifiers actually work, it canreplace other more complicated linearization techniques with simpler andmore effective implementations.

[0048] The models of the non-linear parasitics, non-linear model 1 andnon-linear model 2, may be realized as frequency-independent,memory-less models, using simple look-up tables based on empiricalmeasurements on real parasitics to emulate the non-linear behavior ofthe output parasitics. This approach is perfectly reasonable when theoutput parasitics are mainly in the form of (non-linear) conductances orwhen the bandwidth is relatively small. In most cases, thefrequency-dependent non-linearities in the output can be minimized bycareful separation of the linear and non-linear parts in the model. Ifthis is not sufficient, frequency-dependent adaptive models are used.Any suitable adaptive algorithm such as the well-known LMS (Least MeanSqaure) algorithm or the RLS (Recursive Least Square) algorithm can beused to adjust the non-linear models so that the error component at theoutput is minimized. Frequency-dependent models can be adapted in thesame way, keeping in mind that these models include a memory component,and can generally be seen as non-linear or linear filters.

[0049] Although there may be several output parasitics that affect thelinearity of the output, it is possible to use a model of a singleparasitic to emulate the non-linear amplifier behavior at a given node.For example, all non-linear parasitics at an output node may be groupedinto a single complex-valued parasitic. Alternatively, only the dominantoutput parasitic is modeled. The parasitics usually encountered inpractice are non-linear conductances and capacitances, but the techniqueis not restricted to these parasitics. In fact, the technique is noteven restricted to parasitics, but can be used to compensate for anynon-linear amplifier behavior that can be modeled as a non-linearparasitic. An important example is the compression behavior due tosaturation, which largely can be described by a non-linear drain-sourceimpedance when it comes to its effect on the output.

[0050] It is furthermore important to understand that the proposedcompensation technique does not necessarily have to be used in all thebranches of the composite amplifier, but can be applied to only a singlebranch to compensate for the non-linear parasitics encountered in thesub-amplifier of that branch. This is particularly useful if theparasitics of a specific transistor are much more dominant in generatingdistortion than the parasitics of the other transistor(s) in thecomposite amplifier. The cost for implementing the technique can then begreatly reduced, while most of the distortion is still corrected for.

[0051] A careful analysis of Doherty-type composite amplifiers has alsorevealed that the non-linear output current of the so-called auxiliarysub-amplifier not only generates a desired voltage at the mainsub-amplifier, but also ends up as distortion in the output of thecomposite amplifier, mainly due to reflection of the non-linear outputcurrent at the impedance inverter. The impedance inverter provides aphase shift of 90 degrees only at a single so-called center frequencyand has a growing (chiefly reactive in the lossless case) impedance atfrequencies away from the center frequency. In practice, there willalways be a reflection of the non-linear current from the auxiliaryamplifier at the impedance inverter since the linear output networkimpedance z₂₂ (see FIG. 4) has a strongly frequency-dependent reactivepart in a practical realization. This means that the output will bedistorted at frequencies away from the center frequency. This distortionis present even if all components are linear and lossless, since it isdue to the reflection of the non-linear output current of the auxiliaryamplifier at the impedance inverter. The resulting voltage shows up as astrongly frequency-dependent non-linear component in the amplifiedoutput signal.

[0052] Losses in the transistors, impedance inverters and the DC feednetworks also give rise to unexpected distortion. This is because theselosses make the impedance at the impedance inverter, as seen from theauxiliary amplifier, resistive instead of the ideal short-circuit (alossless quarter wavelength transmission line loaded with the infiniteimpedance of a current generator is a short-circuit at centerfrequency). The distortion in the output caused by these losses are dueto the same type of reflection (but now resistive instead of reactive)of the non-linear current from the auxiliary amplifier at the impedanceinverter which caused the frequency-dependent distortion mentionedabove.

[0053] In many cases, especially for wide-band linear applications, thisdistortion is much more severe than the distortion caused by non-linearparasitics, and therefore it is highly recommendable to try tocompensate for this type of distortion as well. The invention proposes alinear technique for compensating for the distortion that the non-linearoutput current of the auxiliary amplifier causes at the output.Preferably, the non-linear output current from the auxiliary amplifieris emulated based on a linear output network model and merged into theinput signal of the main sub-amplifier, thus effectively compensatingfor distortion caused by the reflection of the non-linear output currentat the impedance inverter.

[0054]FIG. 6 is a schematic block diagram of a composite power amplifierwith compensation for parasitics in combination with compensation forthe non-linear output current of the auxiliary amplifier according to anillustrative embodiment of the invention. The power amplifier 600 shownin FIG. 6 is similar to that of FIG. 5, except for an additional filternetwork 690 for linear compensation.

[0055] In the present solution, the parasitics at each transistor outputnode are preferably separated into two parts, a linear part and aresidual non-linear part. All non-linear parts of the parasitics at anode are conveniently grouped into a single complex-valued non-linearparasitic. The non-linear part of the parasitics at each node arecompensated for in the same manner as described above by means of thefilter network 660 in combination with the models 670, 675 of thenon-linear parts of the parasitics and the adder elements 672, 677. Therest of the output network of the composite amplifier, including thelinear part of the parasitics, is modeled and used for calculatinglinear compensations in the additional filter network 690.

[0056] In a reduced realization, the filter network 690 is provided inthe form of a single filter block H₁₂ which is a cross-coupling filterthat compensates for the non-linear output current of the auxiliaryamplifier.

[0057] Referring once again to the linear output network model of FIG.4, the non-zero impedance z₂₂ will reflect any current i₂ from theauxiliary amplifier as a voltage, and this voltage will be found in theoutput. If the current i₂ was a linear representation of the desiredsignal, this would not be a problem. However, in Doherty and similaramplifiers, this current is a very non-linear function of the desiredsignal due to input drive function f2(x). The non-ideal impedance z₂₂thus makes the amplifier output non-linear.

[0058] By cross-coupling a copy of this non-linear signal (i₂ filteredby impedance z₂₂) to the main amplifier in anti-phase, the distortion atthe output will be cancelled effectively. Since the transimpedance z₂₁is the main linear channel from the main amplifier to the output, thecompensation to the input of main amplifier will linearly transform intoa cancellation signal in the output slightly filtered by z₂₁. Thefiltering effect of the transimpedance z₂₁ should therefore becompensated for in the cross-coupled compensation signal for everythingto cancel perfectly.

[0059] Thus, the cross-coupling filter H₁₂ in a reduced realization ofthe filter network 690 of FIG. 6 may be represented by:

H ₁₂ =z ₂₂ *z ₂₁ ⁻¹   (9)

[0060] where “*” denotes multiplication in the frequency domain orconvolution in the time domain. As can be seen in FIG. 6, thecross-coupled signal is then subtracted from the input signal to mainamplifier.

[0061] Alternatively, instead of cross-coupling the non-linear drivefunction f2(x) through a cross-coupling filter H₁₂, the same effect canbe accomplished by duplicating the non-linear drive function f2(x) inthe upper branch to the main amplifier, using the same filter block H₁₂for compensation.

[0062] In a more elaborate realization, the filter network 690 alsoincludes equalizing filter blocks H₁₁ and H₂₂ as well as an equalizingfilter function in the filter block H₁₂ for providing an equalizedfrequency response. Since the primary function of the auxiliaryamplifier in a Doherty amplifier is to keep the voltage at the mainamplifier below saturation, the frequency dependency of all signals atthe output of main amplifier should be as flat (equalized) as possible.

[0063] In the following, reference will frequently be made to currents,impedances and transimpedances originating from the linear outputnetwork model of FIG. 4, and the following description aims at providingan understanding of how these quantities can be used for providingoptional frequency equalization in the composite amplifier of FIG. 6.

[0064] For the linear component (which constitutes all of i₁ of anuncompensated amplifier) equalization is achieved by means of an inputfilter with the frequency characteristics of z₁₁ ⁻¹, i.e. the inversefilter of the impedance seen at the output of main amplifier.

[0065] For the non-linear component due to i₂, which is filtered throughthe transimpedance z₁₂, and the non-linear part of i₁ that representsthe cross-coupled distortion-canceling signal, which is filtered by z₁₁,the total should have a flat frequency characteristic (not just inmagnitude, but also in phase). Since the non-linear component is formedby two parts, which are differently filtered, and the requirement fordistortion-cancellation at the output dictates a certain relationshipbetween the frequency characteristics of these signals, they should bothbe additionally filtered by the inverse of a special composite filter.Assuming that the raw non-linear function f2(x) has been filtered byz₂₂*z₂₁ ⁻¹ in H₁₂ for the cross-coupled part of i₁ and by nothing forthe auxiliary amplifier part (except for gain), the total compositenon-linear part is represented by: $\begin{matrix}{{\underset{\underset{i_{2}\quad {part}}{}}{{{f2}(x)}*z_{12}} - \underset{{cross}\text{-}{coupled}\quad {part}}{\underset{}{{{f2}(x)}*z_{22}*z_{21}^{- 1}*z_{11}}}} = {{{f2}(x)}*\underset{{composite}\quad {part}}{\underset{}{\left( {z_{12} - {z_{22}*z_{21}^{- 1}*z_{11}}} \right)}}}} & (10)\end{matrix}$

[0066] Thus, the extra equalizing filtering to these signals should havea frequency response defined as the inverse of the composite part:

(z ₁₂ −z ₂₂ *z ₂₁ ⁻¹ *z ₁₁)⁻¹   (11)

[0067] So far, nothing has been said about the magnitudes of thecurrents and voltages in the system except for their relation to eachother. For the lossless case and at (near) the center frequency of thequarter-wave line, the traditional Doherty equations suffice. Forextracting the most power from the chosen transistors, at least one ofthe transistors should be operating at its maximum current I_(max). Thevoltages at peak power should also be the maximum allowed voltageV_(max) (possibly with a safety margin). For a class B amplifier, theoptimal load R_(opt) is V_(max)/I_(max). For an ideal Doherty amplifierthe optimal load impedance depends on the transition point α, such thatR_(o)=R_(opt)(1−α).

[0068] For transition points α below 0.5, the current i₁ should in theideal lossless, narrow-band case vary linearly with the signal amplitudeand be equal to I_(max)(1−α) at the peak amplitude. Current i₂ shouldinstead be zero for output voltages below the transition point, andabove the transition point vary as the (normalized) amplitude minus αdivided by (1−α). This means that auxiliary amplifier delivers currentI_(max) at peak amplitude. For transition points above 0.5 (which isvery unlikely for optimized multi-carrier cases), i₁ would insteadamount to I_(max) at peak amplitude, and i₂ would maximally beI_(max)(1−α)/α.

[0069] The procedure for the lossy, wide-band case is more complicated.The limitations for the currents and voltages are the same as for thenarrow-band lossless case, but the statistical nature of the wide-bandsignals makes it hard to obtain analytical expressions for them. Thevoltages will then depend on the bandwidth used, the amplitudedistribution and phase relations of the individual carriers of thesignal. The lossy, narrow-band case can however provide a startingpoint, from where adjustments can be made for the specific signalsencountered.

[0070] In the lossy case the filter for obtaining the linear part of i₁,as applied to the dimensionless input signal x, will be V_(max)/α*z₁₁⁻¹. The physical meaning of this filter is to generate the current i₁such that the voltage at the output of the current generator of the mainamplifier reaches V_(max) at the normalized input amplitude α when theimpedance seen by this current generator is z₁₁. When observed in thefrequency domain, the term z₁₁ ⁻¹ (the inverse filter of the impedance z₁₁) is equal to 1/z₁₁.

[0071] The filters applied to the non-linear function f2(x) also havethe dimension of current. In practice this is achieved by generating theappropriate drive voltage to the transistors, which act astransconductances, so that the end result is the desired current output.In the lossless case without frequency compensation, the filter appliedto f2(x) for obtaining i₂ is simply a multiplication by j*I_(max) (90degrees phase shift). The maximum amplitude of the function f2(x) ishere assumed equal to one. The cancellation term is then f2(x) filteredby −j*I_(max)*z₂₂*z₂₁ ⁻¹. The compensation (z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)⁻¹ forachieving a frequency-independent non-linear voltage at the mainamplifier can be multiplied to these two expressions in normalized anddimensionless form.

[0072] The expression for obtaining the linear part of i₁ alreadycompensates for losses. The expressions for the non-linear parts must bemodified to do so. Since the relation between the two non-linearcurrents is already established, this is achieved by modifying themagnitude (gain) of both parts equally, so that the amplitude of thesuppression voltage at the main amplifier has the same slope as thelinear parn The factor to multiply with is V_(max)/α divided by(z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)*j*I_(max)/(1−α). The numerator and denominator arethe voltage rise per normalized amplitude for the voltage at the mainamplifier due to the linear part of i₁ and the non-linear currents,respectively. The denominator represents the voltage rise when thecurrent magnitude derived for the narrow-band, lossless case is used.One thing to note here is that the compensation (z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)⁻¹for achieving a frequency-independent non-linear voltage at the mainamplifier is automatically included in this “new” compensation. Thus, inhindsight the normalization is actually not necessary.

[0073] The analytical expressions for obtaining i₂ and i₁ are thus:$\begin{matrix}\begin{matrix}{i_{2} = {\frac{V_{\max}\left( {1 - \alpha} \right)}{\alpha}\left( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} \right)^{- 1}*{{f2}(x)}}} \\{i_{1,{{nonlinear}\quad {part}}} = {{- \frac{V_{\max}\left( {1 - \alpha} \right)}{\alpha}}\underset{{Equalizing}\quad {part}}{\underset{}{\left( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} \right)^{- 1}}}*\underset{\underset{{cancelling}\quad {part}}{{Distortion}\text{-}}}{\underset{}{z_{22}*z_{21}^{- 1}*{{f2}(x)}}}}} \\{i_{1,{{linear}\quad {part}}} = {\frac{V_{\max}}{\alpha}z_{11}^{- 1}*x}}\end{matrix} & \left( {12\text{-}14} \right)\end{matrix}$

[0074] As previously, if the dimensionless signals f2(x) and x arerepresented in the time domain, “*” represent convolution in the timedomain. If they are represented in the frequency domain, the symbolinstead represents multiplication of frequency responses, and themultiplication with inverse filters can be written as a division by thefilter instead. The j and −j factors have vanished from the expressions,but in reality the phases of the currents are about the same as before.The imaginary units are now embedded into the (z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)⁻¹factors. Since z₁₂ (the largest part of the expression, at least nearthe center frequency) mainly represents the transformation of a currentinto a voltage over a quarter-wave line, this entails a 90° phase shiftat the center frequency.

[0075] This means that the filter blocks H₁₁, H₁₂ and H₂₂ in the filternetwork 690 of FIG. 6 may be represented by: $\begin{matrix}\begin{matrix}{H_{11} = {\frac{V_{\max}}{\alpha}z_{11}^{- 1}}} \\{H_{12} = {\frac{V_{\max}\left( {1 - \alpha} \right)}{\alpha}\left( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} \right)^{- 1}*z_{22}*z_{21}^{- 1}}} \\{H_{22} = {\frac{V_{\max}\left( {1 - \alpha} \right)}{\alpha}\left( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} \right)^{- 1}}}\end{matrix} & \left( {15\text{-}17} \right)\end{matrix}$

[0076] In the realization of FIG. 6, the linearly compensated outputsignals of the filter network 690 are used as input to the filternetwork 660. The filter network 660 co-operates with the non-linearmodels 670, 675 for generating the compensation currents required tocompensate for the non-linear part of the parasitics. These compensationcurrents are finally merged with delayed versions of the linearlycompensated output signals of the filter network 690 in order to cancelthe effects of the non-linear part of the parasitics. In this way,large-scale Doherty-specific distortion components such as thenon-linear output current of the auxiliary amplifier is linearlycompensated based on a linear model of the output network, while theresidual distortion caused by the non-linear part of the parasitics iscompensated based on respective non-linear models of the parasitics. Theresult will be a composite amplifier with excellent linearity andoptimized efficiency.

[0077] The proposed linearization techniques effectively remove thetraditional trade-off between linearity and efficiency of Doherty-typeamplifiers, since they can simultaneously optimize linearity andefficiency. Furthermore, they are capable of optimizing linearity andefficiency over large bandwidths with retained performance. Thepossibility of wider relative bandwidths and higher efficiency enablesthe use of composite amplifiers in previously unattainable areas. Forexample, the wider relative bandwidths makes it possible to use theDoherty technique for radio systems at lower frequency, or to makehigh-efficiency amplifiers for entire system bandwidths instead ofsmaller chunks or individual channels. Even if a smaller range ofbandwidth is actually used, the invention enables the making of aunified amplifier with flexible placement of the used bandwidth orchannel within a much larger bandwidth. This implies a lowermanufacturing cost, since fewer variants have to be manufactured.

[0078] Sometimes it can be useful to design reduced variants of theproposed composite amplifier, with simplified filter networkconfigurations. For example, when the voltage at the output node of themain amplifier has an equalized frequency response due to the proposedequalization, the linear output network model can be simplified to theextent that the ideal voltage at the output node of the main amplifiercan be described by a simple combination of the input signal x andf2(x). In practice, this means that the filters G₁₁ and G₁₂ (models ofthe impedance z₁₁ and the transimpedance z₁₂) are not needed whencalculating the ideal output node voltages. In addition, in all caseswhere the non-linear drive function f2(x) should not be seen at theoutput of the auxiliary amplifier, this a priori knowledge can also beused to reduce the number of filters. All paths that go from f2(x) tothe input of non-linear model of the lower branch can actually beremoved. These two ideas can be combined as illustrated in FIG. 7, withsignificantly reduced complexity as a result.

[0079]FIG. 7 is a schematic block diagram of a favorable, simplifiedimplementation of a composite amplifier according to the currently mostpreferred embodiment of the invention. The composite amplifier 700comprises a main amplifier 710, an auxiliary amplifier 720, an antennaload 730, an impedance inverter 740, an input drive arrangement 750 forthe auxiliary amplifier, suitable non-linear models of parasitics 770,775, a delay block 780 and a filter network 790. As can be seen, thefilter network 790 has a much simpler configuration than the combinationof the filter networks 660 and 690 in the composite amplifier of FIG. 6.The filter blocks H₁₁, H₁₂ and H₂₂ are defined according to expressions(15-17) to provide cancellation of distortion and to provide anequalized frequency response. Due to the equalized frequency response ofthe output of the main amplifier, the ideal output node voltage of themain amplifier can be determined as a simple combination of the inputsignal x and f2(x) by using the adder element 782.

[0080] The resulting signal is delayed by the delay block 780 tosynchronize the non-filtered path to the filtered paths, andsubsequently applied to the non-linear model 770 to generate theappropriate compensation current. The compensation current is finallymerged with the linearly compensated signal from the filter network 790in the adder element 772 to generate an equalized and fully compensatedinput signal to the main amplifier 710. The convolution of the filterblocks H₁₁ and G₂₁ is now treated as a single filter H₁₁*G₂₁, whichdetermines the ideal output node voltage of the auxiliary amplifier inresponse to the input signal x. The determined ideal output node voltageis then applied to the non-linear model 775 for generating theappropriate compensation current. This compensation current is finallymerged with the linearly compensated signal from the filter block H₂₂ inthe adder element 777 to provide an equalized and fully compensatedinput signal to the auxiliary amplifier 720. The example illustrated inFIG. 7 shows that, for some of the most important cases, a specificimplementation can be very simplified compared to the more generalsolution of FIG. 6.

[0081] Although the filter networks described above with reference toFIGS. 5-7 may seem complicated, since they are assembled from manyfrequency-dependent impedances and transimpedances, the filtercomplexity can be reduced in several ways. In a digital implementation,the filters can be assembled from measured impedances by multiplicationand division in the frequency domain. The thereby assembled filters canthen either be used directly for filtering in the frequency domain, orbe converted to time-domain filters. A frequency-domain window can beapplied for restricting the filters to suitable bandwidths. Typically,filters are implemented as FIR (Finite Impulse Response) filters forflexibility.

[0082] In practice, the performance of the described techniques willdepend on how well the characteristics of the Doherty output network areknown. Measuring trans-impedances in the output network is often hard todo directly, since the (RF) voltage probe and the current injector willalways have parasitics that must be taken into account. Indirectly,impedance parameters (Z-parameters) can be extracted by traveling wavemeasurements (S-parameters). A combination of different parameters thatare easy to measure can also be selected. The required filters oremulating networks can then be designed using extracted impedances andtransimpedances.

[0083] Many different implementations are possible. For example, digitalor analog signal processing can be used, and the processing can beperformed with a variety of techniques, at base-band, intermediate orfinal (RF) frequencies. Arbitrary combinations of these can be used,matching the requirements for a function with a convenient way ofimplementing it. The solutions can be used statically, optimized at thetime of manufacture or at specific times during maintenance, ordynamically adaptive, for continuously optimizing the linearity andefficiency of the amplifier.

[0084] The invention can be used with the non-linear parts of theparasitics separated from an otherwise linear network as describedabove, or with the entire parasitics (including the linear as well asnon-linear parts) separated. A further variety of this idea is toseparate all conductive parasitics from the rest of the output network.

[0085] Although the invention has been described with reference to atwo-stage Doherty-type composite amplifier, it is evident that theinvention is applicable to composite amplifiers with more than twostages as well as to other types of composite amplifiers that share someor all of the described characteristics.

[0086] The embodiments described above are merely given as examples, andit should be understood that the present invention is not limitedthereto. Further modifications, changes and improvements which retainthe basic underlying principles disclosed and claimed herein are withinthe scope and spirit of the invention.

References

[0087] [1] F. H. Raab, “Efficiency of Doherty RF Power AmplifierSystems”, IEEE Trans. Broadcasting, vol. BC-33, no. 3, pp. 77-83,September 1987.

[0088] [2] U.S. Pat. No. 5,420,541.

[0089] [3] U.S. Pat. No. 5,568,086.

[0090] [4] U.S. Pat. No. 5,786,727.

[0091] [5] U.S. Pat. No. 5,025,225.

[0092] [6] D. M. Upton et al. “A New Circuit Topology to Realize HighEfficiency, High Linearity, and High Power Microwave Amplifiers”, IEEEProc. RAWCON'98.

[0093] [7] The International Patent Application WO 97/20385.

[0094] [8] U.S. Pat. No. 4,999,585.

What is claimed is:
 1. A linearization method for a composite amplifier(500; 600; 700) having at least two sub-amplifiers (510, 520; 610, 620;710, 720), said method comprising the steps of: for at least onesub-amplifier: determining a sub-amplifier output signal based on alinear model of the output network of the composite amplifier; andemulating and compensating for, in the input signal to thesub-amplifier, a non-linear amplifier behavior based on a non-linearmodel of a sub-amplifier parasitic using the determined sub-amplifieroutput signal as input to the model.
 2. The linearization methodaccording to claim 1, wherein said emulating step includes the step ofdetermining a compensation signal according to the non-linear model byapplying the determined sub-amplifier output signal as input to thenon-linear model; and said compensating step includes the step ofmerging the compensation signal into the sub-amplifier input signal tocompensate for said non-linear amplifier behavior.
 3. The linearizationmethod according to claim 1, wherein said step of determining asub-amplifier output signal is based at least partly on an interactionbetween two sub-amplifiers.
 4. The linearization method according toclaim 3, wherein said interaction is represented by a transimpedancebetween the two sub-amplifiers.
 5. The linearization method according toclaim 1, wherein said non-linear amplifier behavior is the compressionbehavior due to amplifier saturation, said compression behavior beingmodeled as a non-linear sub-amplifier parasitic.
 6. The linearizationmethod according to claim 1, wherein said steps of determining asub-amplifier output signal and emulating and compensating for anon-linear amplifier behavior are performed for each sub-amplifier (510,520; 610, 620; 710, 720).
 7. The linearization method according to claim1, wherein said method further comprises steps of emulating andcompensating for, in the input signal to a main sub-amplifier (510; 610;710), the non-linear output signal of an auxiliary sub-amplifier (520;620; 720) based on a linear model of the output network of the compositeamplifier.
 8. The linearization method according to claim 1, whereinsaid method further comprises the steps of equalizing, for at least oneof said sub-amplifiers, the amplifier frequency response.
 9. A compositeamplifier (500; 600; 700) having at least two sub-amplifiers (510, 520;610, 620; 710, 720), wherein said composite amplifier comprises: for atleast one sub-amplifier: means (560; 660; 782, H₁₁*G₂₁) for determininga sub-amplifier output signal based on a linear model of the outputnetwork of the composite amplifier; means (570, 572, 575, 577; 670, 672,675, 677; 770, 772, 775, 777) for emulating and compensating for, in theinput signal to the sub-amplifier, a non-linear amplifier behavior basedon a non-linear model of a sub-amplifier parasitic using the determinedsub-amplifier output signal as input to the model.
 10. The compositeamplifier according to claim 9, wherein said emulating means includesmeans (570, 575; 670, 675; 770, 775) for determining a compensation0signal according to the non-linear model in response to the determinedsub-amplifier output signal as input to the non-linear model; and saidcompensating means includes means (572, 577; 672, 677; 772, 777) formerging the compensation signal into the sub-amplifier input signal tocompensate for said non-linear amplifier behavior.
 11. The compositeamplifier according to claim 9, wherein said means for determining asub-amplifier output signal is based at least partly on an interactionbetween two sub-amplifiers.
 12. The composite amplifier according toclaim 11, wherein said interaction is represented by a transimpedancebetween the two sub-amplifiers.
 13. The composite amplifier according toclaim 9, wherein said composite amplifier further comprises means (690;790) for emulating and compensating for, in the input signal to a mainsub-amplifier (610; 710), the non-linear output signal of an auxiliarysub-amplifier (620; 720) based on a linear model of the output networkof the composite amplifier.
 14. The composite amplifier according toclaim 9, wherein said composite amplifier further comprises, for atleast one of said sub-amplifiers, means for equalizing the amplifierfrequency response.
 15. A transmitter (200) having a composite poweramplifier (220; 500; 600; 700) based on at least two sub-amplifiers(510, 520; 610, 620; 710, 720), wherein said composite power amplifiercomprises: for at least one sub-amplifier: means (560; 660; 782,H₁₁*G₂₁) for determining a sub-amplifier output signal based on a linearmodel of the output network of the composite amplifier; means (570, 572,575, 577; 670, 672, 675, 677; 770, 772, 775, 777) for emulating andcompensating for, in the input signal to the sub-amplifier, a non-linearamplifier behavior based on a non-linear model of a sub-amplifierparasitic using the determined sub-amplifier output signal as input tothe model.
 16. The transmitter according to claim 15, wherein saidemulating means includes means (570, 575; 670, 675; 770, 775) fordetermining a compensation signal according to the non-linear model inresponse to the determined sub-amplifier output signal as input to thenon-linear model; and said compensating means includes means (572, 577;672, 677; 772, 777) for merging the compensation signal into thesub-amplifier input signal to compensate for said non-linear amplifierbehavior.
 17. The transmitter according to claim 15, wherein said meansfor determining, for at least one sub-amplifier, a sub-amplifier outputsignal is based at least partly on an interaction between twosub-amplifiers.
 18. The transmitter according to claim 17, wherein saidinteraction is represented by a transimpedance between the twosub-amplifiers.
 19. The transmitter according to claim 15, wherein saidcomposite power amplifier further comprises means (690; 790) foremulating and compensating for, in the input signal to a mainsub-amplifier (610; 710), the non-linear output signal of an auxiliarysub-amplifier (620; 720) based on a linear model of the output networkof the composite amplifier.
 20. The transmitter according to claim 15,wherein said composite power amplifier further comprises means forequalizing, for at least one of said sub-amplifiers, the amplifierfrequency response.